I have a raspberry and an auxiliary PCB with transistors for driving some LED strips.
The strips datasheets says 12V, 13.3W/m, i'll use 3 strips in parallel, 1.8m each, so 13.3*1.8*3 = 71,82W, with 12 V, almost 6A.
I'm using an 8A transistor, E13007-2.
In the project i have 5 channels of different LEDs: RGB and 2 types of white.
R, G, B, W1 and W2 are directly connected in py pins.
LED strips are connected with 12V and in CN3, CN4 for GND (by the transistor).
Transistor schematic.
I know that that's a lot of current passing through the transistors, but, is there a way to reduce the heating? I think it's getting 70-100°C. I already had a problem with one raspberry, and i think it's getting dangerous for the application. I have some large traces in the PCB, that's not the problem.
Some thoughts:
1 - Resistor driving the base of the transistor. Maybe it won't reduce heating, but i think it's advisable for short circuit protection, how can i calculate this?
2 - The PWM has a frequency of 100Hz, is there any difference if i reduce this frequency?
The BJT transistor you're using has current gain hFE of roughly 20. This means that the collector current is roughly 20 times the base current, or the base current needs to be 1/20 of the collector current, i.e. 6A/20=300mA.
Rasperry PI for sure can't supply 300mA current from the IO pins, so you're operating the transistor in linear region, which causes it to dissipate a lot of heat.
Change your transistors to MOSFETs with low enough threshold voltage (like 2.0V to have enough conduction at 3.3V IO voltage) to keep it simple.
Using a N-Channel MOSFET will run much cooler if you get enough gate voltage to force to completely enhance. Since this is not a high volume item why not simply use a MOSFET gate driver chip. Then you can use a low RDS on device. Another device is the siemons BTS660 (S50085B BTS50085B TO-220). it is a high side driver that you will need to drive with an open collector or drain device. It will switch 5A at room temperature with no heat sink.It is rated for much more current and is available in a To220 type package. It is obsolete but available as is the replacement. MOSFETs are voltage controlled while transistors are current controlled.
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I am working with an Ettus N310 that is being controlled by some 3rd party software. I don't have much of an insight of how they set up and control the device, just tell it what center frequency to tune to and when to grab IQ. If I receive a signal, let's say a tone, at or very near the center frequency, I end up with a large DC offset that jumps around every few 100 usec. If I offset the signal well away from the center frequency, the DC offset is negligible. From what I see in Ettus' documentation, DC offset compensation is something that's set once when the device starts receiving but it looks to me like here it is being done periodically while the USRP is acquiring data. If I receive a signal near center frequency, the DC offset compensator gets messed up and creates a worse bias. Is this a feature on the N310 that I am not aware of or is this probably something that the 3rd party controller is doing?
Yes, there's a DC offset compensation in the N310. The N310 uses an Analog Devices RFIC (the AD9371), which has these calibrations built-in. Both the AD9371 and the AD9361 (used in the USRP E3xx and B2xx series) don't like narrow-band signals close to DC due to their calibration algorithms (those chips are optimized for telecoms signals).
Like you said, the RX DC offset compensation is happening at initialization. At runtime, the quadrature error correction kicks in. The manual holds a table of those: https://uhd.readthedocs.io/en/latest/page_usrp_n3xx.html#n3xx_mg_calibrations). You can try turning off the QEC tracking and see if it improves your system's performance.
When I record a signal with USRP N320 SDR, it has some problems on the edges of the spectrum. For example, when I choose sample rate 50 Msps, 2 MHz of the start of the spectrum and 2 MHz of the end of the spectrum, gives the wrong results. When it see a pulse on the edges it decreases the power and changes the frequency little bit. But 46 MHz bandwidth is perfectly working.
Sample rate: 50 Msps, Properly working bandwidth: 46 MHz
Sample rate: 100 Msps, Properly working bandwidth: 90 MHz
Sample rate: 200 Msps, Properly working bandwidth: 180 MHz
I tried to filter the edges with bandpass filter but it does give the OOOOOO problem. Even if I choose the sample rate 50 Msps. But normally, I can record successfully without bandpass filter when I choose sample rate 200 Msps.
Is there a solution to record the edges correctly. Or filtering it without dropping samples.
First off:
I tried to filter the edges with bandpass filter but it does give the OOOOOO problem
means that your computer isn't fast enough to apply the filter to the data stream. That might mean two things: you've designed a filter that's too long and could be shorter and still do what you want, or what you want to do requires a filter of that length and you will need to find a faster PC (hard) or use a faster filter implementation (did you try the FFT filters?).
For example, when I choose sample rate 50 Msps, 2 MHz of the start of the spectrum and 2 MHz of the end of the spectrum, gives the wrong results.
This is not surprising! Remember that anything with a ADC needs an anti-aliasing filter on the analog side, and these can't be arbitrarily sharp. So, necessarily, the spectrum at the edge of your band gets a bit dampened, and there's a bit of aliasing there. The dampening, you could counteract by throwing an equalizing filter on your PC at it, which would need to necessarily be more compute-intense than what is happening on the USRP, but the aliasing of the lowest frequencies onto the highest, and vice versa, due to finite steepness of the analog anti-aliasing filter you cannot repair. That's the signal processing blues for any kind of acquisition device.
There's one trick though, which the USRP uses: when your requested sampling rate is lower than the ADC's sampling rate, the USRP can internally apply a (better!) digital filter to select that target sampling rate as bandwidth, and decimate to that.
Thus, depending on the ADC rate to output sampling rate relationship (in UHD, the ADC rate is called "master clock rate", MCR), there's further digital filtering and decimation going on in the digital logic inside the N320. These filters also can't be infinitely sharp – and you might see that.
Generally, you'd want that decimation between MCR and the sampling rate you've requested to be an even number, and not too large. Don't have the N320's digital signal processing architecture in my head right now, but I bet using a decimation that's a multiple of 4 or even 8 is a good move – you get to use the nicer half-band filters then.
Modern UHD also has the filter API, with which you can work with these digital filters manually; this rarely is what you really want to do here, though.
I'm using STM32F411 with USB CDC library, and max speed for this library is ~1Mb/s.
I'm creating a project where I have 8 microphones connected into ADC line (this part works fine), I need a 16-bit signal, so I'm increasing accuracy by adding first 16 signals from one line (ADC gives only 12-bits signal). In my project, I need 96k 16-bit samples for one line, so it's 0,768M signals for all 8 lines. This signal needs 12000Kb space, but STM32 have only 128Kb SRAM, so I decided to send about 120 with 100Kb data in one second.
The conclusion is I need ~11,72Mb/s to send this.
The problem is that I'm unable to do that because CDC USB limited me to ~1Mb/s.
Question is how to increase USB speed to 12Mb/s for STM32F4. I need some prompt or library.
Or maybe should I set up "audio device" in CubeMX?
If small b means byte in your question, the answer is: it is not possible as your micro has FS USB which max speeds is 12M bits per second.
If it means bits your 1Mb (bit) speed assumption is wrong. But you will not reach the 12M bit payload transfer.
You may try to write (only if b means bit) your own class but I afraid you will not find a ready made library. You will need also to write the device driver on the host computer
Is the internal clock on the ATTiny85 sufficiently accurate for one-wire timing?
Per https://learn.sparkfun.com/tutorials/ws2812-breakout-hookup-guide one-wire timing seems to need accuracy around the 0.05us range, so a 10% clock error on the AVR at 8MHZ would cause 0.0125us sized timing differences (assuming the 10% error figure is accurate, and that it's 10% error on frequency, not +/- 10% variance on each pulse).
Not a ton of margin - but is it good enough?
First of all, WS2812 LEDs are not the 1-wire.
The control protocol of WS2812 is described in the datasheet
The short answer is yes, ATTiny85, also the whole AVR family have enough clock accuracy to control the WS2812 chain. But routine should be written at assembler, also no interrupts should be allowed, to guarantee match the timing requests. When doing the programming well, 8MHz speed of the internal oscillator may be enough to output the different data to two WS2812 chains simultaneously.
So, when running 8MHz ±10%, the one clock cycle would be 112...138 ns.
The datasheet requires (with 150ns tolerance):
When transmitting "one": high level to be 550...850ns; - 6 clock cycles (672...828) matches this range (also 5 clock cycles (560...690ns) matches)
following low level - 450...750ns; - 5 cycles (560...690ns)
When transmitting "zero": high level 200...500ns; - 3 cycles (336...414ns)
following low level 650...950ns; - 6 cycles (672...828).
So, as you can see, considering tolerance ±10% of the clock's source, you can find the integer number of cycles which will guarantee match to the required intervals.
Speaking from the experience, it still be working if the low level, which follows the pulse, will be extended for a couple hundreds of nanoseconds.
There are known issues using internal oscilator with UART - should be timed to 2% accuracy while the internal oscilator can be up to 10% off with factory setting. While it can be calibrated(AVR has register OSCCAL for that purpose), its frequency is influenced by temperature.
It is worth the try, but might not to be reliable with temperature changes or fluctuating operating voltage.
References: ATmega's internal oscillator - how bad is it, Timing accuracy on tiny2313, Tuning internal oscilator
The timing requirements of NeoPixels (WS2812B) are wide enough that the only really critical part is the minimum width of a 1 bit. The ATtiny85 at 16Mhz is plenty fast to drive a string of them from a GPIO pin. At 8Mhz, it may not work (I haven't tried yet). I just released a small Arduino sketch which allows you to control NeoPixel strings of any length on a ATtiny85 without using any RAM.
https://github.com/bitbank2/NeoPixel
For devices with hardware SPI (e.g. ATMega328p), it's better to use SPI to shift out the bits (also included in my code).
I have been using FRDM_KL46Z development board to do some IR communication experiment. Right now, I got two PWM outputs with same setting (50% duty cycle, 38 kHz) had different voltage levels. When both were idle, one was 1.56V, but another was 3.30V. When the outputs were used to power the same IR emitter, the voltages were changed to 1.13V and 2.29V.
And why couldn't I use one PWM output to power two IR emitters at the same time? When I tried to do this, it seemed that the frequency was changed, so two IR receivers could not work.
I am not an expert in freescale, but how are you controlling your pwm? I'm guessing each pwm comes from a separate timer, maybe they are set up differently. Like one is in 16 bit mode (the 3.3V) and the other in 32 (1.56v) in that case even if they have the same limit in the counter ((2^17 - 1) / 2) would be 50% duty cycle of a 16 bit timer. But in a 32 bit, that same value would only be 25% duty so, one output would be ~1/2 the voltage of the other. SO I suggest checking the timer setup.
The reason the voltage changed is because the IR emmiters were loading the circuit. In an ideal situation this wouldn't happen, but if a source is giving too much current the voltage usually drops a bit.